Few people know what it is Mosfet, but almost everyone has heard that it is very good. Let's first understand this word. MOSFET- English abbreviation for metal-oxide-semiconductor field effect transistor. Its structure consists of a metal and a semiconductor, separated by a layer of silicon dioxide (SiO2). In general, the structure is called MIS (metal-dielectric-semiconductor).
Transistors based on such structures, unlike bipolar ones, are controlled by voltage rather than current and are called unipolar transistors, since their operation requires the presence of charge carriers of only one type. High temperature stability, low control power, low susceptibility to breakdown, self-limiting drain current, high speed in switching mode, low noise level - these are the main advantages of MOSFET field-effect transistors over radio tubes and bipolar transistors.
Most lovers of high-quality sound reproduction rate an amplifier based on MOSFET transistors as very high level, almost like tube amplifiers, because compared to conventional amplifiers bipolar transistors They produce a softer sound, create less distortion and are resistant to overload. MOSFETs are superior to classic tube amplifiers, both in damping coefficient and in transmitting low and high frequencies. The cutoff frequency of such amplifiers is significantly higher than that of an amplifier based on bipolar transistors, which has a beneficial effect on the sound.
Powerful MOSFET field-effect transistors have a smaller spread of basic parameters than bipolar transistors, which makes their parallel connection easier and reduces the total output resistance of the power amplifier.
Simple MOSFET amplifier circuit
Amplifier parameters
- Output Power (RMS): 140 W into 8 ohms, 200 W into 4 ohms.
- frequency range: 20 Hz - 80 kHz -1dB.
- Input sensitivity: 800 mV at a power of 200 W at 4 ohms.
- Distortion:<0.1% (20 Гц - 20 кГц).
- Signal to Noise Ratio: >102dB unweighted, 105dB (A-weighted considering 200W into 4 ohms).
The figure shows a circuit of one of the simplest UMZCHs using field-effect transistors of this type in the output stage. And its power is as much as 200 watts! This MOSFET power amplifier is suitable for many purposes such as high-power concert guitar or home theater. The amplifier has a good frequency range - from 1 dB 20 Hz to 80 kHz. Distortion rate less than 0.1% at full power, and the signal-to-noise ratio is better than -100 dB. Further simplification is possible through the use of an op-amp in the preamplifier stage.
The entire ULF structure is housed in a small aluminum housing. The circuit is powered by a simple bipolar rectifier with a 250-watt toroidal transformer. Please note that the photo shows a monoblock - that is, a single-channel amplifier, since it is assembled for an electric guitar.
The radiator is made of black anodized aluminum profile. The case is 300 mm long and is equipped with a rear 80 mm cooling fan. The fan runs constantly, so the radiator is always cool, even at maximum power (or at least slightly above ambient temperature).
Old but golden
Old but goldenAmplifier circuitry has already gone through a spiral in its development and now we are witnessing a “tube renaissance”. In accordance with the laws of dialectics that were so persistently drummed into us, a “transistor renaissance” should follow. The very fact of this is inevitable, because lamps, for all their beauty, are very inconvenient. Even at home. But transistor amplifiers have their own shortcomings...
The reason for the “transistor” sound was explained back in the mid-70s - deep feedback. It gives rise to two problems at once. The first is transient intermodulation distortion (TIM distortion) in the amplifier itself, caused by signal delay in the feedback loop. There is only one way to combat this - by increasing the speed and gain of the original amplifier (without feedback), which can seriously complicate the circuit. The result is difficult to predict: either it will happen or not.
The second problem is that deep feedback greatly reduces the output impedance of the amplifier. And for most loudspeakers this is fraught with the occurrence of those same intermodulation distortions directly in the dynamic heads. The reason is that when the coil moves in the gap of the magnetic system, its inductance changes significantly, so the impedance of the head also changes. With a low output impedance of the amplifier, this leads to additional changes in the current through the coil, which gives rise to unpleasant overtones, mistakenly taken for distortion of the amplifier. This can also explain the paradoxical fact that with an arbitrary choice of speakers and amplifiers, one set “sounds” and the other “does not sound.”
secret of tube sound =
high output impedance amplifier
+ shallow feedback
.
However, similar results can be achieved with transistor amplifiers. All the circuits given below have one thing in common - an unconventional and now forgotten “asymmetrical” and “irregular” circuit design. However, is she as bad as she is made out to be? For example, a bass reflex with a transformer is a real Hi-End! (Fig. 1) And the phase inverter with a divided load (Fig. 2) is borrowed from tube circuitry...
Fig.1
Fig.2
Fig.3
These schemes are now undeservedly forgotten. But in vain. Based on them, using a modern element base, it is possible to create simple amplifiers with very high quality sound. In any case, what I collected and listened to sounded decent - soft and “tasty”. Depth feedback in all circuits it is small, there is local feedback, and the output impedance is significant. There is no general environmental protection for direct current.
However, the given diagrams work in the classroom B, therefore they are characterized by “switching” distortions. To eliminate them, it is necessary to operate the output stage in a “pure” class A. And such a scheme also appeared. The author of the scheme is J.L.Linsley Hood. The first mentions in domestic sources date back to the second half of the 70s. 
Fig.4
The main disadvantage of class amplifiers A, limiting the scope of their application is a large quiescent current. However, there is another way to eliminate switching distortions - the use of germanium transistors. Their advantage is low distortion in mode B. (Someday I will write a saga dedicated to germanium.) Another question is that these transistors are not easy to find now, and the choice is limited. When repeating the following designs, you need to remember that the thermal stability of germanium transistors is low, so there is no need to skimp on radiators for the output stage.
 Fig.5 | This diagram shows an interesting symbiosis of germanium transistors with field effect transistors. The sound quality, despite the more than modest characteristics, is very good. To refresh the impressions from a quarter of a century ago, I took the time to assemble the structure on a mock-up, slightly modernizing it to suit modern part values. Transistor MP37 can be replaced with silicon KT315, since during setup you will still have to select the resistance of resistor R1. When operating with an 8 Ohm load, the power will increase to approximately 3.5 W, the capacitance of capacitor C3 will have to be increased to 1000 μF. And to work with a 4 Ohm load, you will have to reduce the supply voltage to 15 volts so as not to exceed maximum power scattering of output stage transistors. Since there is no overall DC OOS, the thermal stability is only sufficient for home use. |
Fig.6
Fig.7 Author: A.I. Shikhatov (drafting and comments) 1999-2000
Published: collection "Designs and diagrams for reading with a soldering iron" M. Solon-R, 2001, pp. 19-26.
- Schemes 1,2,3,5 were published in the magazine "Radio".
- Scheme 4 is borrowed from the collection
V.A. Vasiliev "Foreign amateur radio designs" M. Radio and Communications, 1982, pp. 14...16 - Schemes 6 and 7 are borrowed from the collection
J. Bozdekh "Design of additional devices for tape recorders" (translated from Czech) M. Energoizdat 1981, p. 148,175 - Details about the mechanism of intermodulation distortion: Should the UMZCH have a low output impedance?
UMZCH on field effect transistors
UMZCH on field-effect transistorsThe use of field-effect transistors in a power amplifier can significantly improve sound quality while simplifying the overall circuit. The transfer characteristic of field-effect transistors is close to linear or quadratic, so there are practically no even harmonics in the spectrum of the output signal; in addition, the amplitude of higher harmonics quickly decreases (as in tube amplifiers). This makes it possible to use shallow negative feedback in field-effect transistor amplifiers or to abandon it altogether. After conquering the vastness of “home” Hi-Fi, field-effect transistors began to attack car audio. The published diagrams were originally intended for home systems, but maybe someone will risk applying the ideas contained in them in a car...
Fig.1
This scheme is already considered classic. In it, the output stage, operating in AB mode, is made of MOS transistors, and the preliminary stages are of bipolar ones. The amplifier provides fairly high performance, but to further improve the sound quality, bipolar transistors should be completely excluded from the circuit (next picture). 
Fig.2
After all reserves for improving sound quality have been exhausted, only one thing remains - a single-ended output stage in “pure” class A. The current consumed by the preliminary stages from a higher voltage source in both this and the previous circuit is minimal. 
Fig.3
The output stage with a transformer is a complete analogue of tube circuits. This is for a snack... The integrated current source CR039 sets the operating mode of the output stage. 
Fig.4
However, a wideband output transformer is a rather complex unit to manufacture. An elegant solution - a current source in the drain circuit - was proposed by the company
The use of field-effect transistors in the input stages of low-frequency amplifiers designed to operate from high-impedance signal sources makes it possible to improve the transmission coefficient and significantly reduce the noise figure of such amplifiers. The high input impedance of the PT avoids the need to use large capacitance capacitors. The use of PT in the first stage of a ULF radio receiver increases the input impedance to 1-5 MOhm. Such a ULF will not load the final stage of the intermediate frequency amplifier. Using this property of field-effect transistors (high R input), a number of circuits can be significantly simplified; At the same time, dimensions, weight and energy consumption from the power source are reduced.
This chapter discusses the principles of construction and circuits of ULF on field-effect transistors with a p-n junction.
The field-effect transistor can be connected in a circuit with a common source, a common drain and a common gate. Each of the switching circuits has certain characteristics on which their application depends.
COMMON SOURCE AMPLIFIER
This is the most commonly used DC circuit and is characterized by high input impedance, high output impedance, voltage gain greater than unity, and signal inversion.
In Fig. 10a shows a circuit of a common-source amplifier in which there are two power supplies. The signal voltage generator Uin is connected to the input of the amplifier, and the output signal is taken between the drain and the common electrode.
A fixed bias is disadvantageous, since it requires an additional power source, and is generally undesirable for the reason that the characteristics of a field-effect transistor vary significantly depending on temperature and have a large scatter from instance to instance. For these reasons, in most practical circuits with field-effect transistors, an automatic bias is used, created by the current of the field-effect transistor itself across the resistor R and (Fig. 10, b) and similar to the automatic bias in tube circuits.
Rice. 10. Schemes for connecting PTs with a common source.
a - with a fixed offset; b - with automatic shift; c - with zero offset; g - equivalent circuit.
Let's consider a circuit with zero bias (Fig. 10, c). At sufficiently low frequencies, when the resistance of capacitors C z.s (Fig. 10, d) and C z.i can be neglected in comparison with R z, the voltage gain can be written:
(1)
where R i is the dynamic resistance of the PT; it is defined as follows:
Here we note that SR i = μ, where μ is the transistor’s own voltage gain.
Expression (1) can be written differently:
(2)
In this case, the output impedance of the amplifier (Fig. 10, c)
(3)
With automatic displacement (Fig. 10, b), the cascade mode is determined by a system of equations:
The solution of this system gives the value of the drain current I s at the DC operating point:
(4)
At set value I c from expression (4) we find the resistance value in the source circuit:
(5)
If the voltage value U s.i. is specified, then
(6)
The slope value for a cascade with automatic bias can be found by the expression
(7)
COMMON DRAIN AMPLIFIER
A cascade with a common drain (Fig. 11, a) is often called a source follower. In this circuit, the input impedance is higher than in a common source circuit. The output impedance here is low; There is no inversion of the signal from input to output. The voltage gain is always less than unity, and nonlinear signal distortion is insignificant. The power gain can be large due to the significant ratio of input and output impedances.
A source follower is used to obtain a small input capacitance, to convert the impedance downward, or to handle a large input signal.
Rice. 11. Circuits of amplifiers with a common drain.
a - the simplest source follower; b - equivalent circuit; c - source follower with increased bias resistance.
At frequencies where 1/ωСз.и is significantly greater than R i and R n (Fig. 11, b), the input and output voltages are related to each other by the relation
where is the voltage gain K and
(8)
Where 
The input impedance of the cascade shown in Fig. 11, a, is determined by the resistance R z. If R 3 is connected to the source, as shown in Fig. 11, V, the input impedance of the amplifier increases sharply:
(9)
So, for example, if R з = 2 MΩ, and the voltage gain K and = 0.8, then the input resistance of the source follower is 10 MΩ.
The source follower input capacitance for a purely ohmic load is reduced due to the inherent feedback of this circuit:
The output resistance R out of the source follower is determined by the formula
(11)
When R i >>R n, which often occurs in practice, according to (11) we have:
(12)
For high load resistances
Rout ≈ 1/S (13)
Source follower output capacitance
(4)
It must be said that the gain of the source follower weakly depends on the amplitude of the input signal, and therefore this circuit can be used to work with a large input signal.
COMMON GATE AMPLIFIER
This switching circuit is used to convert low input impedance to high output impedance. The input resistance here has approximately the same value as the output resistance in a circuit with a common drain. A common gate cascade is also used in high-frequency circuits, since in most cases there is no need to neutralize internal feedback.
Voltage gain for common gate circuit
(15)
where R r is the internal resistance of the input signal generator.
Stage input impedance
(16)
and the day off
(17)
SELECTION OF PT OPERATING POINT
The choice of the operating point of the transistor is determined by the maximum output voltage, maximum power dissipation, maximum change in drain current, maximum voltage gain, the presence of bias voltages, and minimum noise figure.
To achieve the maximum output voltage, you must first select the highest supply voltage, the value of which is limited by the permissible drain voltage of the transistor. To find the load resistance at which the maximum undistorted output voltage is obtained, we define the latter as the half-difference between the power supply voltage E p and the saturation voltage (equal to the cutoff voltage). Dividing this voltage by the selected value of the drain current at the operating point I s, we obtain the optimal value of the load resistance:
(18)
The minimum value of power dissipation is achieved at minimum voltage and drain current. This parameter is important for portable equipment powered by batteries. In cases where the requirement for minimum power dissipation is of paramount importance, it is necessary to use transistors with low voltage cut-off U ots. Drain current can be reduced by changing the gate bias voltage, but one must be aware of the reduction in transconductance that accompanies decreasing drain current.
The minimum temperature drift of the drain current for some transistors can be achieved by aligning the operating point with the point on the transistor flow characteristic that has a zero temperature coefficient. In this case, for the sake of accurate compensation, the interchangeability of transistors is sacrificed.
The maximum gain at low load resistance values is achieved when the transistor operates at the point with maximum transconductance. For field-effect transistors with a control p-n junction, this maximum occurs when the gate-source voltage is zero.
The minimum noise figure is achieved by establishing a low voltage regime at the gate and drain.
SELECTION OF FIELD TRANSISTOR BY CUT-OFF VOLTAGE
In some cases, the choice of DC cutoff voltage has a decisive influence on the operation of the circuit. Transistors with low cutoff voltage have a number of advantages in circuits that use low-power power supplies and where greater temperature stability is required.
Consider what happens when two FETs with different cutoff voltages are used in a common-source circuit at the same supply voltage and zero gate bias.
Rice. 12. Characteristics of PT transmission.
Let us denote U ots1 - cut-off voltage of transistor PT1 and U ots2 - cut-off voltage of transistor PT2, while U ots1
U c1 =U c2 =U c ≥U ots2
Let's introduce the term “quality indicator”:
(20)
The value of M can be understood from Fig. 12, which shows a typical transmission characteristic of a p-channel FET.
The slope of the curve at U z.i =0 is equal to S max. If the tangent at the point U z.i =0 is continued until it intersects with the abscissa axis, then it will cut off the segment U ots /M on this axis. This is easy to show based on (20):
(21)
Consequently, M is a measure of the nonlinearity of the flow characteristic of a field-effect transistor. B shows that when manufacturing field-effect transistors using the diffusion method, M = 2.
Let's find the value of the current I c0 using expression (21):
Substituting its value into (19), we get:
If in formula (1) we put R i >>R n, then the voltage gain for a circuit with a common source
(23)
Substituting the value of the gain (23) into expression (22), we obtain:
(24)
From relation (24) we can draw the following conclusion: at a given supply voltage, the cascade gain is inversely proportional to the cutoff voltage of the field-effect transistor. Thus, for field-effect transistors manufactured by the diffusion method, M = 2 and at U ots1 = 1.5 V (KP103E), U ots2 = 7 V (KP103M), supply voltage 12.6 V and U c = 7 V, the gain factors of the cascades are equal to 7.5 and 1.6, respectively. The gain of the cascade with PT1 increases even more if, by increasing the load resistance R n, U c is reduced to 1.6 V. It should be noted that in this case, with a constant supply voltage E p, a transistor with a low transconductance can provide a higher voltage gain than a transistor with a higher transconductance (due to higher load resistance).
In the case of low load resistance Rн, it is advisable to use field-effect transistors with a high cut-off voltage to obtain a higher gain (due to an increase in S).
For transistors with a low cut-off voltage, the change in drain current from temperature is much smaller than for transistors with a high cut-off voltage, and therefore the requirements for stabilizing the operating point are lower. At gate biases that set the temperature coefficient of drain current to zero, transistors with a lower cutoff voltage have a higher drain current than a transistor with a higher cutoff voltage. In addition, since the gate bias voltage (at zero temperature coefficient) of the second transistor is higher, the transistor will operate in a mode in which the nonlinearity of its characteristics is more affected.
For a given supply voltage, FETs with a low cutoff voltage allow you to get more dynamic range. For example, from two transistors having a cutoff voltage of 0.8 and 5 V with a supply voltage of 15 V and a maximum load resistance calculated from relation (18), at the output of the first one can obtain double the amplitude of the output signal (defined as the difference between E p and U ots), equal to 14.2 V, while in the second - only 10 V. The difference in gain will be even more obvious if E p is reduced. So, if the supply voltage is reduced to 5 V, then the double amplitude of the output voltage of the first transistor will be 4.2 V, but the second transistor is almost impossible to use for these purposes.
NONLINEAR DISTORTION IN AMPLIFIERS
The amount of nonlinear distortion that occurs in DC amplifiers is determined by many circuit parameters: bias, operating voltage, load resistance, input signal level, characteristics of field-effect transistors.
When a sinusoidal voltage U 1 sinωt is applied to the input of an amplifier with a common source, the instantaneous value of the total voltage in the gate-source circuit can be written
U z.i = E cm + U 1 sinωt
where E cm is the external bias voltage applied to the gate.
Taking into account the quadratic dependence of the drain current on the gate voltage (1), the instantaneous value of i c will be equal to:
(24a)
Opening the brackets in equation (24a), we obtain a detailed expression for the drain current:
From expression (24b) it is clear that the output signal, along with the constant component and the first harmonic, contains the second harmonic of the input signal frequency.
Nonlinear distortion is determined by the ratio of the rms value of all harmonics to the rms value of the fundamental harmonic in the output signal. Using this definition, from expression (24b) we find the harmonic coefficient, expressing (E cm -U ots) through I c0:
(24v)
Expression (24c) gives only an approximate result, since the real flow characteristics of the PT differ from those described by expression (1).
To achieve minimal nonlinear distortion it is necessary:
Maintain the value of U c.i large enough so that at the maximum drop in the output signal the condition is met
U s.i ≥(1.5...3)U s.i.
Do not operate at gate-drain voltages close to breakdown;
- choose a load resistance large enough.
In Fig. 16, c shows a circuit in which a field-effect transistor operates with a large R n, which ensures low distortion and high gain. The second field-effect transistor T2 is used here as a load resistance. This circuit provides a voltage gain of about 40 dB at E supply = 9 V.
The choice of the type of FET that provides the least distortion depends on the input signal level, supply voltage and required bandwidth. With a large output signal level and significant bandwidth, a PT with a large Uref is desirable. At a low input signal level or low supply voltage, PTs with low Us are preferable.
GAIN STABILIZATION
The ULF gain on the PT, as on other active elements, is subject to the influence of various destabilizing factors, under the influence of which it changes its value. One of these factors is changes in ambient temperature. To combat these phenomena, the same methods are generally used as in circuits based on bipolar transistors: they use negative feedback in both current and voltage, covering one or more cascades, and introduce temperature-dependent elements into the circuit.
In a field-effect transistor with a p-n junction, under the influence of temperature, the reverse-biased gate current changes exponentially, and the drain current and transconductance change.
The effect of changing the gate current I g on the gain can be weakened by reducing the resistance of the resistor R g in the gate circuit. To reduce the influence of changes in drain current, as in the case of bipolar transistors, negative DC feedback can be used (Fig. 13a).
Let's take a closer look at some ways to reduce the impact of changes in slope S on the gain.
In boost mode weak signals The gain of the uncompensated stage on the field-effect transistor drops as the temperature rises. For example, the gain of the circuit in Fig. 13, a, equal to 13.5 at 20° C, decreases to 12 at +60° C. This decrease is primarily due to the temperature change in the slope of the field-effect transistor. Bias parameters such as drain current I s, gate-source voltage U g.i and source-drain voltage U c.i change slightly due to the existing DC feedback.
Rice. 13. Amplifier circuits with gain stabilization.
a - uncompensated cascade; b - compensated amplification stage; c - compensated amplification stage with OOS; g -transient characteristic.
By including several ordinary diodes in the negative feedback circuit between the gate and source (Fig. 13, b), it is possible to stabilize the gain of the amplifier without introducing additional stages. As the temperature increases, the forward voltage of each diode decreases, which in turn leads to a decrease in voltage U c.i.
It has been experimentally shown that the resulting change in voltage moves the operating point in such a way that the slope S is relatively stable within certain limits of temperature change (Fig. 13, d). For example, the gain of the amplifier according to the circuit in Fig. 13, b, equal to 11, practically retains its value within the range of temperature changes of 20-60 ° C (K and changes by only 1%).
The introduction of negative feedback between gate and source (Fig. 13, c) reduces the gain, but provides better stability. Amplifier gain according to the diagram in Fig. 13, c, equal to 9, practically does not change when the temperature changes from 20 to 60°.
By carefully selecting the operating point and number of diodes, the gain can be stabilized with an accuracy of 1% over a range of up to 100°C.
REDUCING THE INFLUENCE OF PT INPUT CAPACITANCE ON THE FREQUENCY PROPERTIES OF AMPLIFIERS
For the source follower shown in Fig. 11, a, according to its equivalent circuit (Fig. 11, b), the time constant of the input circuit can be determined with sufficient accuracy for practical calculations as follows:
τ in = R g [C g + C z.s + C z.i (1 - K i)], (25)
where R g and C g are the parameters of the signal source.
From expression (25) it is clear that the time constant of the input circuit is directly dependent on the capacitances C z.s and C z.i, and the capacitance Cz.i due to the influence of the environmental protection is reduced by (1-K and) times.
However, obtaining a voltage gain close to unity (in order to eliminate the influence of capacitance C s.i.) in a conventional source follower circuit is fraught with difficulties associated with the low breakdown voltage of the field-effect transistor. So, in order to obtain a voltage gain of 0.98 on a field-effect transistor KP102E with a maximum drain current I c0 = 0.5 mA, a maximum slope of 0.7 mA/V, it is necessary to use a resistance R n = 65 kOhm. At I c0 = 0.5 mA, the voltage drop across the resistance R n will be about 32.5 V, and the supply voltage should be at least greater than this voltage by the amount U ots, i.e. E p = 35 V.
To avoid the need to use a high supply voltage to obtain a gain close to unity, combined follower circuits based on field-effect and bipolar transistors are often used in practice.
In Fig. 14, a shows a combined circuit both by the type of transistors used in it and by the circuit of their connection, called a source follower with a tracking link. The drain of field-effect transistor T1 is connected to the base of bipolar transistor T2, from the collector of which the signal is supplied to the source terminal of the field-effect transistor in antiphase with the input signal. By selecting resistors R5 and R6, the signal voltage at the source can be obtained equal to the input voltage, thereby eliminating the influence of capacitance C.
Resistor R1 installed in the gate bias circuit is connected to the source of transistor T1 through a large capacitance capacitor C2. The effective resistance in the bias circuit is determined by the resistance of the resistor R 1 and the feedback coefficient, so that
(35)
where U and is the signal amplitude at the source of transistor T1.
Rice. 14. Amplifier circuits with reduced input capacitance.
a - source follower with tracking link; b - with reduced capacity C z.s; c - source follower with dynamic load.
For large values of β of bipolar transistor T2, the gain of the circuit can be approximately estimated by the following expression:
(36)
If the amplifier is designed to operate at low frequencies, then resistor R6 can be bypassed with capacitor C3 (shown with a dotted line in Fig. 14a); in this case, the upper frequency limit is determined by the expression
(37)
Above, we discussed a method for reducing the influence of the gate-source capacitance C on the frequency response of the amplifier by obtaining a gain close to unity at the source follower. The influence of capacitance C z.s remained unchanged.
Further improvement in the frequency characteristics of amplifiers can be achieved by weakening the static gate-drain capacitance in the input circuit of the circuit.
To reduce the influence of the capacitance between the gate and the drain, you can use a method similar to that described above to reduce the influence of the capacitance C g.i., i.e., reduce the signal voltage across the capacitor. In the diagram shown in Fig. 14, b, the influence of capacitance C z.s is reduced so much that the input capacitance of the cascade is almost completely determined by the arrangement of parts in the circuit and the installation capacitance.
The first stage on transistor T1 has a small load in the drain circuit and is a source follower for the signal taken from the source. The output signal is fed to a common collector stage that uses a bipolar transistor.
To reduce the influence of capacitance C 3.s, the signal from the output stage (emitter follower) is fed through capacitor C2 to the drain of transistor T1 in phase with the input signal. To increase the compensation effect, it is necessary to take measures to increase the transmission coefficient of the first stage. This is achieved by applying a signal from the emitter follower to the bias resistor R3. As a result, the voltage applied to the drain becomes greater, and the negative feedback becomes more effective. In addition, increasing the transmission coefficient of the first stage further reduces the influence of capacitance C z.i.
If you do not use the listed methods for reducing the gate capacitance, then the input capacitance, as a rule, is quite significant (for the KP103 transistor it is 20-25 pF). As a result, it is possible to reduce the input capacitance to 0.4-1 pF.
A source follower with a dynamic load (Based on materials from Yu. I. Glushkov and V. N. Semenov), covered by tracking feedback to the drain, is shown in Fig. 14, c. With the help of such a circuit, it is possible to eliminate the influence of the static gain of the field-effect transistor μ on the transmission coefficient of the source follower, and also to reduce the capacitance C g.s. Transistor T2 acts as a stable current generator, setting the current in the source circuit of field-effect transistor T1. Transistor T3 is a dynamic load in the drain circuit of a field-effect transistor running on alternating current. Source follower parameters:
ECONOMICAL VLF
The developer is sometimes faced with the task of creating economical low-frequency amplifiers operating from a low-voltage power supply. In such amplifiers, field-effect transistors with low cut-off voltage U otc and saturation current I c0 can be used; these circuits have undoubted advantages over tube and bipolar transistor circuits.
The choice of operating point in economical field-effect transistor amplifiers is determined based on the condition of obtaining minimum power dissipation. To do this, the bias voltage U c.i is selected almost equal to the cut-off voltage, while the drain current tends to zero. This mode ensures minimal heating of the transistor, which leads to low gate leakage currents and high input resistance. The required gain at low drain currents is achieved by increasing the load resistance.
In economical low-frequency amplifiers, the cascade circuit shown in Fig. is widely used. 10, b. In this circuit, a bias voltage is generated across the resistance in the source circuit, which creates a negative current feedback that stabilizes the mode from the influence of temperature fluctuations and parameter variations.
We can propose the following procedure for calculating economical ULF cascades made according to Fig. 10, b.
1. Based on the condition for obtaining minimum power dissipation, we select a field-effect transistor with a low cut-off voltage U ots and saturation current I c0.
2. Select the operating point of the field-effect transistor by current I c (units - tens of microamps).
3. Considering that at a bias voltage close to the cutoff voltage, the drain current can be approximately determined by the expression
Rc ≈ U ots /R and (38)
source circuit resistance
Ri ≈ U ots /I and (39)
4. Based on the required gain, we find R n. Since the gain
(40)
then, neglecting the shunting effect of the differential drain-source resistance R i and substituting instead of S its value obtained by differentiating the expression for the drain current in (40), we obtain:
(41)
From the last expression we find the required load resistance:
(42)
This is where the calculation of the amplifier ends and during the adjustment process the values of resistors R n and R i are only specified.
In Fig. Figure 15 shows a practical diagram of an economical low-frequency amplifier operating from a capacitive sensor (for example, from a piezoceramic hydrophone).
Due to the low bias current of the output amplifier, consisting of two transistors T2 and T3, the power dissipation of the entire preamplifier is 13 μW. The preamplifier consumes 10 µA current at a supply voltage of 1.35 V.
Rice. 15. Schematic diagram of an economical amplifier.
The input impedance of the preamplifier is determined by the resistance of resistor R1. The actual input resistance of the field-effect transistor can be neglected, since it is an order of magnitude greater than the resistance of resistor R1.
In small-signal mode, the preamplifier input stage is equivalent to a common-source circuit, while the bias circuits are designed like a source follower circuit.
The field-effect transistor used in this circuit must have a small cut-off voltage Uots and a small drain current I c0 at a gate voltage U z.i = 0.
The conductivity of the field-effect transistor channel T1 depends on the drain current, and since the latter is insignificant, the conductivity is also small. Therefore, the output resistance of a common-source circuit is determined by the resistance of resistor R2. According to the output impedance of the amplifier is 4 kOhm, the voltage gain is 5 (14 dB).
ULF CASCADES WITH DYNAMIC LOAD
Field-effect transistors make it easy to implement low-frequency amplifier circuits with dynamic loads. Compared to a rheostatic amplification stage, in which the load resistance is constant, an amplifier with a dynamic load has a higher voltage gain.
The schematic diagram of an amplifier with a dynamic load is shown in Fig. 16, a.
An active element, field-effect transistor T2, is used as the dynamic resistance of the drain load of field-effect transistor T1, the internal resistance of which depends on the amplitude of the signal at the drain of transistor T1. Transistor T1 is connected according to a common source circuit, and T2 is connected according to a common drain circuit. For direct current, both transistors are connected in series.
Rice. 16. Schematic diagrams of amplifiers with dynamic loads.
a - on two PTs; b - on a PT and a bipolar transistor; c - with a minimum number of details.
The input signal Uin is supplied to the gate of field-effect transistor T1, and is removed from the source of transistor T2.
The amplification cascade (Fig. 16, a) can serve as a standard one when constructing multistage amplifiers. When using field-effect transistors of the KP103Zh type, the cascade has the following parameters:
It should be noted that when using FETs with a low cut-off voltage, it is possible to obtain a higher voltage gain than when using FETs with a high cut-off voltage. This is explained by the fact that a PT with a low cut-off voltage has a higher internal (dynamic) resistance than a PT with a high cut-off voltage.
An ordinary bipolar transistor can also be used as dynamic resistance. In this case, the voltage gain is even slightly higher than when using a field-effect transistor in a dynamic load (due to the larger R i). But in this case, the number of parts required to build an amplification stage with a dynamic load increases. The schematic diagram of such a cascade is shown in Fig. 16, b, and its parameters are close to the parameters of the previous amplifier shown in Fig. 16, a.
Amplifiers with dynamic loads should be used to obtain high gain in low-noise ULFs with low supply voltage.
In Fig. 16, c shows an amplification stage with a dynamic load, in which the number of parts is reduced to a minimum, and this circuit provides a gain of up to 40 dB with a low noise level. The voltage gain for this circuit can be expressed by the formula
(43)
where S max1 is the transconductance of transistor T1; R i1, R i2 are the dynamic resistances of transistors T1 and T2, respectively.
ULF ON CHIPS
The K2UE841 type microcircuit is one of the first linear microcircuits mastered by our industry. It is a two-stage amplifier with deep negative feedback (follower), assembled using field-effect transistors. Microcircuits of this type are widely used as input stages of sensitive broadband amplifiers, as remote stages when transmitting signals through a cable, in active filter circuits and other circuits that require high input and low output impedance and a stable transmission coefficient.
The circuit diagram of such an amplifier is shown in Fig. 17, a; methods for switching on the microcircuit are shown in Fig. 17, b, c, d.
Resistor R3 is included in the circuit to protect the output transistor from overloads in the event of short circuits at the output. By slightly reducing the feedback (in Fig. 17, in R oс shown by a dotted line), you can obtain a transmission coefficient equal to unity or slightly more.
The input resistance of repeaters can be significantly increased (10-100 times) if feedback is provided to the gate circuit through capacitor C (shown by the dotted line in Fig. 17, c). In this case, the input impedance of the repeater is approximately equal to:
R in = R h / (1-K u),
where K and is the repeater transmission coefficient.
The basic electrical parameters of the repeater are as follows:
The industry has mastered the production of hybrid film microcircuits of the K226 series, which are low-noise low-frequency amplifiers with a field-effect transistor at the input. Their main purpose is to amplify weak signals alternating current from sensors with high internal resistance.
Rice. 17. Chip K24E841.
a - schematic diagram; b - circuit with one power supply with a voltage of 12.6 V; c - circuit with two power supplies with a voltage of +-6.3 V; d - circuit with one power supply with a voltage of -6.3 V.
The microcircuits are made on a glass-ceramic substrate using hybrid film technology using field-effect and bipolar transistors.
Low-frequency amplifier microcircuits are divided into groups according to gain and noise level (Table 1). Appearance And dimensions are presented in Fig. 18.
Fundamental electrical circuits amplifiers are shown in Fig. 19, a, b and 20, a, b, and their connection diagrams are shown in Fig. 21, a, d. When switching on the microcircuits according to the diagrams in Fig. 21, a and b, the input resistance of the amplifiers is equal to the resistance of the external resistor R i. To increase the input resistance (up to 30 MOhm or more), it is necessary to use the circuits in Fig. 21.6, g.
| Chip types | Gain | Noise voltage, µV |
| K2US261A | 300 | 5 |
| K2US265A | 100 | 5 |
| K2US261B | 300 | 12 |
| K2US265B | 100 | 12 |
| K2US262A | 30 | 5 |
| K2US262B | 30 | 12 |
| K2US263A | 300 | 6 |
| K2US263B | 300 | 12 |
| K2US264A | 10 | 6 |
| K2US264B | 10 | 12 |
Table 1
Rice. 18. Appearance and overall dimensions of K2US261-K2US265 microcircuits.
Basic electrical parameters of K2US261 and K2US262 microcircuits:
| Supply voltage | +12.6 V +-10% -6.8V +-10% |
| Power consumption: | |
| from +12.6 V source | No more than 40 mW |
| from -6.3 V source | No more than 50 mW |
| Changing the gain in the operating temperature range (from -45 to +55°C) | +-10% |
| Intrinsic noise voltage in the 20 Hz - 20 kHz band depending on the groups (with the input shorted with a 5000 pF capacitor) | 5 µV and 12 µV |
| 3 MOhm | |
| Output impedance | 100 Ohm |
| Input capacitance | 15 pF |
| Upper limit frequency at level 0.7 | Not less than 200 kHz |
| Lower cutoff frequency | Determined by the external filter capacities |
| The maximum output voltage on an external load is 3 kOhm in a frequency band up to 100 kHz with a nonlinear distortion coefficient of no more than 5% | Not less than 1.5 V |
Rice. 19. Schematic diagrams of amplifiers.
a - K2US261; b - K2US262.
Rice. 20. Schematic diagrams of amplifiers.
a - K2US263; b - K2US264 (all diodes are KD910B type).
Basic electrical parameters of K2US263 and K2US264 microcircuits:
| Supply voltage | +6 V ±10% -9 V +-10% |
| Power consumption: | |
| from +6 V source | 10 mW |
| from source - 9 V | 50 mW (K2US263), 25 mW (K2US264) |
| Change in gain in the operating temperature range (from -45 to +55° C) | +-10% |
| Input impedance at 100 Hz | Not less than 10 MOhm |
| Input capacitance | No more than 15 pF |
| Output impedance | 100 Ohm (K2US263), 300 Ohm (K2US264) |
| Upper limit frequency with an output signal amplitude of at least 2.5 V and frequency response unevenness + -5% | 100 kHz (K2US263), 200 kHz (K2US264) |
| Lower cutoff frequency | Determined by the external filter capacity |
| THD at 2.5 V output voltage | 5% (K2US263), 10% (K2US264) |
Rice. 21. Amplifier connection circuits.
Recommendations for the use of microcircuits. The frequency dependence and cut-off frequency at the level of 0.7 V in the low-frequency region with a sufficiently large time constant of the input circuit is determined by the external negative feedback filter capacitor C2 and the resistance of the feedback circuit resistor R o.c in accordance with the relations:
Peak voltages at the input of microcircuits K2US261, K2US262 should not exceed 1 V for positive polarity and 3 V for negative; at the input of the K2US263, K.2US264 microcircuits - no more than 2 V for positive polarity and no more than 1 V for negative polarity.
Leakage resistance R1 for input current in the operating temperature range -60 to +70° C should not exceed 3 MOhm. At lower maximum temperatures or when output voltage requirements are reduced, the value of resistor R1 can be increased to increase the input impedance of the stage.
The leakage current of the input decoupling capacitor C1 should not exceed 0.06 μA.
To maintain the maximum output voltage, the leakage current of capacitor C2 in the operating temperature range should not exceed 20 μA. This requirement is satisfied by a K52-1A capacitor with a capacity of 470 μF, the leakage current of which does not exceed 10 μA at these voltages.
PRACTICAL DIAGRAMS OF LOW FREQUENCY AMPLIFIERS USING FIELD TRANSISTORS
Typically, field-effect transistors are used in amplifiers in conjunction with bipolar transistors, but they can also be used as active devices in multistage amplifiers audio frequency with resistive-capacitive coupling. In Fig. Figure 22 shows an example of using field-effect transistors in an RC amplifier circuit. The circuit of this amplifier was used for recording sound signals seas. The signal to the input of the amplifier was taken from the piezoceramic hydrophone G, and the load of the amplifier was a cable of the KVD4x1.5 type, 500 m long.
The input stage of the amplifier is made of a field-effect transistor type KP103Zh with minimal noise figure. For the same purpose (reducing noise), the first two stages are powered by a reduced voltage obtained using the D1R8 parametric stabilizer. Thanks to these measures, the noise level referred to the input in the frequency band 4 Hz-20 kHz was 1.5-2 μV.
To adjust the frequency response of the amplifier in the area higher frequencies In parallel with resistors R6 and R10, you can connect the corresponding correction capacitors.
To match the high output resistance of the amplifier with a low-resistance load (cable), a voltage follower on transistors T4, T5 is used, which is a two-stage amplifier with direct coupling. To eliminate the shunting effect of bias resistors R11, R12, positive feedback on alternating current is introduced through the chain R13, C6. The calculated value of the output resistance of such a repeater is 10 Ohms.
To check the functionality and gain of the amplifier, use a calibration generator assembled using a symmetrical multivibrator circuit. The calibration generator produces rectangular amplitude-stabilized pulses with a frequency of 85 Hz using D2-D5 zener diodes of type D808, which, when the calibrator is turned on, are supplied through a hydrophone to the input of the amplifier. Using a voltage divider on resistors R16, R17, the pulse amplitude was set to 1 mV.
Despite the simplicity of the amplifier circuit, the gain changes slightly (about 2%) when the ambient temperature changes in the range of 0-40 ° C, and the gain at room temperature 20 ° C was equal to 150.
Rice. 22. Schematic diagram of a hydroacoustic amplifier.
If the output resistance of the first stage on a field-effect transistor can be reduced so much that it becomes possible to use conventional bipolar transistors in subsequent stages, then it is not economical to use field-effect transistors for further amplification. In these cases, amplifiers using field-effect and bipolar transistors are used.
In Fig. Figure 23 shows a schematic diagram of a low-frequency amplifier using field-effect and bipolar transistors, which has parameters close to those of a three-stage RC amplifier using field-effect transistors (Fig. 22). So, with a gain equal to 150, frequency response at level 0.7 from 20 Hz to 100 kHz, the value of the maximum output undistorted signal at R n = 3 kOhm is 2 V.
Field-effect transistor T1 (Fig. 23) is connected according to a circuit with a common source, and the bipolar transistor is connected according to a circuit with a common emitter. To stabilize performance characteristics, the amplifier is covered by negative DC feedback.
In Fig. Figure 24 shows the circuit of a low-frequency amplifier with direct couplings, developed by V.N. Semenov and V.G. Fedorin, designed to amplify weak signals from sources with high input impedance. The amplifier does not contain coupling capacitors, so its dimensions can be small.
The amplifier parameters are as follows:
The circuit is a UPT with 100% DC feedback; Due to this, a minimum of drift and stability of the regimes are achieved. The DC feedback is introduced through a low-pass filter, so the lower cut-off frequency of the amplifier is determined by the parameters of this filter.
To stabilize the gain, negative feedback is used at the signal frequency with a depth of about 20 dB. The gain depends on the depth of feedback.
Rice. 23. Schematic diagram of ULF on field-effect and bipolar transistors.
Rice. 24. Schematic diagram of a ULF with direct connections.
The use of feedback makes the amplifier uncritical to changes in supply voltage and variations in the parameters of transistors and all parts except R10 and R11. Features of the circuit include the fact that transistors T3 and T4 operate with voltages U b.e. equal to U b.e.
The high input impedance of the amplifier is achieved through the use of field-effect transistors. At lower frequencies it will be determined by the resistance of resistor R1, at upper frequencies - by the input capacitance of the circuit.
A.G. Milekhin
Literature:
- Field effect transistors. Physics, technology and application. Per. from English edited by A. Mayorova. M., " Soviet radio", 1971.
- Sevin L. Field-effect transistors. M., “Soviet Radio”, 1968.
- Malin V.V., Sonin M.S. Parameters and properties of field-effect transistors. M., "Energy", 1967.
- Sherwin V. Causes of distortion in field-effect transistor amplifiers. - "Electronics"‚ 1966, No. 25.
- Downs R. Economical preamplifier. "Electronics", 1972, No. 5.
- Holzman N. Elimination of emissions using an operational amplifier. "Electronics", 1971, No. 3.
- Gozling V. Application of field-effect transistors. M., "Energy". 1970.
- De Kold. The use of diodes for temperature stabilization of the gain of a field-effect transistor - "Electronics", 1971, No. 12.
- Galperin M.V., Zlobin Yu.V., Pavleiko V.A. DC transistor amplifiers. M., "Energy", 1972.
- Technical catalogue. “New devices. Field effect transistors. hybrid integrated circuits" Ed. Central Research Institute "Electronics", 74.
- Topchilov N. A. Hybrid linear microcircuits with high-resistance input - “Electronic Industry”, 1973, No. 9.
If sound volume is not the most important thing, but preference is given to sound quality, then this UMZCH will come in handy. The output stage, made according to a push-pull circuit on a complementary pair of powerful field-effect transistors with an insulated gate, provides sound quality subjectively akin to “tube”.
Yes, the objective characteristics are not bad at all:
Sound amplifier based on field-effect transistors
The low frequency pre part is done on A1. The signal from its output is fed to a push-pull output stage using opposite field-effect transistors with an insulated gate - 2SK1530 (n-channel) and 2SJ201 (p-channel). The required bias voltage is created at the gates of the transistors using resistors R8, R9 and diodes VD3 and VD4.
Diodes eliminate “step” distortion by creating an initial potential difference between the gates of the field-effect transistors. The stabilizing voltage of the OOS is removed from the output of the output stage and, through the circuit R4-C6, is supplied to the inverse input of the operational amplifier A1, which is also the input.
The voltage gain depends on the ratio of the resistances of resistors R1 and R4. By changing the resistance R1, you can adjust the sensitivity of this UMZCh within a fairly wide range, adapting it to the output parameters of the existing preliminary UMZCH. However, you should know that, as usual, increasing sensitivity leads to increased distortion. So there has to be a reasonable compromise here.
The supply voltage is ±25V; you can use an unstabilized source, but it must be well filtered from AC background ripples. The operational amplifier is powered by a bipolar voltage of ±18V from two parametric stabilizers based on zener diodes VD1 and VD2. Instead of the 2SK1530 transistor, you can use the older 2SK135, 2SK134. Instead of the 2SJ201 transistor, you can use 2SJ49, 2SJ50.
Transistors must be installed on a heat sink. Transistors 2SK1530 and 2SJ201 have such a housing design that they do not have a radiator plate in contact with the crystal; their housing is made of ceramic plastic, which conducts heat well, but does not conduct electricity. Therefore, transistors can be installed on a common radiator. If transistors with radiator plates that have electrical contact with the crystal are used, then it is necessary to install them on different radiators, isolated from each other, or use careful insulation using mica spacers.
In any case, there must be a heat-conducting paste between the heat-removing surface of the transistor body and the radiator; it covers irregularities in the contact between the transistor body and the radiator and thus increases the actual contact area, which contributes to better heat dissipation. The audio operational amplifier can be replaced with almost any op-amp, for example, or some other option. 1N4148 diodes can be replaced with KD522 or KD521.
The 1N4705 zener diodes can be replaced with any other zener diodes designed for a stabilization voltage of 18V, or each of them can be replaced with two zener diodes connected in series, giving a total of 18V (for example, 9V and 9V). Capacitors C1 and C4 must be for a voltage of at least 35V, capacitors C7 and C8 for a voltage of at least 50V. Despite the presence electrolytic capacitors C7 and C8 for power supply, at the output of the power source there must be capacitors of a significantly larger capacity to ensure high-quality suppression of AC ripple at the output of the power source.
The installation is made on a printed circuit board made of foil fiberglass with a one-sided arrangement of printed tracks (Fig. 2). Preparation method printed circuit board can be any available. The printed tracks do not have to be exactly the same shape as those shown in the figure - it is important that the necessary connections are provided.
Below are schematic diagrams and articles on the topic “ULF on field-effect transistors” on the radio electronics and radio hobby site.
What is “ULF on field-effect transistors” and where is it applied, schematic diagrams of homemade devices that relate to the term “ULF on field-effect transistors”.
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